1. Field of the Invention
The present invention relates to the field of instrumentation amplifiers.
2. Prior Art
Instrumentation amplifiers are often used to sense small differential voltages having a common mode voltage near the negative or positive supply-rail voltages. Instrumentation amplifiers with a so-called three operational amplifier (opamp) architecture are not able to sense in a range close to the rail unless separate level shifts are used at the input. More specifically, these architectures use voltage feedback around operational amplifiers to the input, and the feedback output voltage cannot go below the negative rail or above the positive rail without a level shift. Moreover, the voltage feedback around the opamp reduces its common mode rejection ratio.
There are two major options. Firstly, the rail sensing can be obtained with switched or ‘flying’ capacitors [Ref. 1: LTC6800 Spec. sheet]. This has the disadvantage of a relative low bandwidth. Secondly, the rail sensing can be achieved by a continuous-time current feedback instrumentation amplifier [Ref. 2: Bernard van den Dool], [Ref. 3: U.S. Pat. No. 6,559,720]. A general block diagram for such an amplifier is shown in FIG. 1.
Instrumentation amplifiers with current feedback for sensing at the rail voltage conventionally use simple voltage-to-current (V-I) converters, as shown in FIG. 3 [Ref. 2: Berhard van den Dool]. The non-linearity of these simple V-I converters cancels when used in an instrumentation amplifier as shown in FIG. 1. The relative high offset, and poor common-mode rejection ratio (CMRR), can be largely improved by the chopper instrumentation amplifier architecture of FIG. 2. But residual inaccuracies and non-linearities of the simple differential amplifier stages of the order of 0.1% are still too much for accurate applications. Therefore, ways to improve the accuracy are desired.
Voltage feedback or voltage boosting can be applied around the P-channel input transistors to improve their accuracy, as shown in FIG. 4. Voltage feedback reduces the influence of the input transistors on the accuracy and linearity. The result is that the transconductance G of the input stages is fully determined by the source resistors Rs, so that G becomes equal to G=2/(RS1+RS2). These resistors can be chosen as both accurate and linear. However the ability to sense around the negative supply rail is lost again in the architecture of FIG. 4, because the output voltages of the operational amplifiers cannot be drawn below the negative supply rail voltage.